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Although many modern narrowband FM (NBFM) transmitters use a phase locked loop (PLL) system to directly generate RF output at the transmitter output frequency, the use of traditional crystal oscillator and multiplier chain still has some merits. Among these are possibly better phase noise characteristics, freedom from microphonics, good stability, simple circuitry, and relative freedom from the danger of obsolescence caused by manufacturer discontinuance of the LSI IC. This IC is the heart of the PLL circuit and generally is a sole source component. Many service technicians are more comfortable with the traditional oscillator-multiplier approach, which has been used for many years. The use of generic parts makes servicing easier, especially in areas where parts procurement is difficult. On the other hand, the more modern approach using a PLL is simpler, easier to tune up, and more versatile, and the better choice for current product design (as long as the LSI IC continues to be available). An example of each approach is shown.
The first transmitter to be described is a VHF NBFM transmitter that employs the traditional oscillator-multiplier approach. The transmitter is intended for use on the 2-meter (144-148 MHz) amateur band. This transmitter uses a crystal oscillator that employs a varactor diode and inductor in series with the crystal. A fundamental mode crystal operating in the 18-MHz range-one eighth the output frequency-is used. The voltage on the varactor is modulated with the audio to be transmitted. This is effectively a narrowband voltage-controlled crystal oscillator (VCXO). The output circuit of the crystal oscillator contains a double-tuned circuit to pass the second harmonic of the crystal at 36.6 MHz. The output of the crystal oscillator feeds a frequency doubler and is multiplied by two, producing output at 73.3 MHz. This is followed by another doubler that produces the final output frequency at 146.52 MHz, and then a power amplifier stage delivering 0.25 watts to a 50-ohm load. This is adequate for short- and medium-range transmission up to 1-2 miles with a simple "rubber ducky" or quarter-wave whip antenna. It is possible to extend this range to several miles by working through a repeater, as is the usual practice in 2-meter (146-MHz) amateur radio work. A crystal cut for 18.315 MHz is used, and this gives output on 146.520 MHz, the commonly used frequency for simplex work. This circuit could be operated anywhere between 130-180 MHz with appropriate component substitutions. A fundamental cut crystal is much better regarding its ability to be used in a VCXO than is a third overtone crystal, with better modulation characteristics, because the effective Q of a fundamental crystal is often somewhat lower than a third overtone crystal and less tricky to "pull" in operating frequency, exactly what we want in a VCXO. The extra multiplier stage acts as a buffer, which may be needed with the third overtone oscillator approach.
Referring to the schematic of the RF section shown in FIG. 1, crystal oscillator Q1 is a Colpitts type with the crystal acting as the inductor, C1 and C2 the feedback capacitors, and L11 and the varactor diode as an auxiliary LC network in series with the crystal. L11 is used to set the transmitter on the exact frequency and has a range of a few kilohertz. A fundamental cut crystal cut for 20 pf capacitance parallel resonant should be used. R1, R2, and R3 are bias resistors for Q1. In the collector circuit of Q1 is a double-tuned circuit tuned to the second harmonic of the crystal, L1, C4, C5, L2, C6, and C7 serving as a 36-MHz bandpass filter. R4 and C3 are decoupling and bypass components for the oscillator stage. The second harmonic energy is fed to the base of doubler Q2. R5 is a bias resistor. The collector circuit of Q2 consists of another double-tuned circuit nominally tuned to 73 MHz. C8, L4, C9, L5, C10, and C11 make up this network. R6 and C12 are decoupling and bypass components for the first doubler stage. Q3 is another doubler similar to the Q2 stage except for component values. Trimmers C14, C16, coupling capacitor C15, and L6 and L7 are tuned to 146 MHz, with R8 and C13 serving as decoupling and bypass components. The next stage Q4 is a power amplifier stage that delivers about 0.25- 0.3 watt output to a 50-ohm load with a 7.5-volt supply.
FIG. 1 Schematic of a 0.25 to 0.5 W Narrowband FM VHF Transmitter (144-148
MHz)
With a 12-volt supply, as much as 1 watt can be obtained. L8, C17, L9, C18, L10, and C19 are used for matching the impedance of the power output stage to a 50-ohm load and to reduce harmonic components to less than 40 dBc. This stage runs at 50- 60 percent efficiency, and the oscillator-multiplier chain Q1-Q2-Q3 draws about 30 mA at 7.5 volts. The entire RF section draws about 100 mA at 7.5 volts. IC1 is used to derive a regulated 5-volt bias for the varactor diode and the audio amplifier stage to be discussed later (see FIG. 2). R10 and R11 feed bias to the varactor diode, and audio is fed through R11 and coupling capacitor C21. The audio voltage modulates the bias on the varactor diode, varying the crystal oscillator frequency, and therefore producing an FM signal as the transmitter output. Deviation is set to 5 kHz by means of the audio level applied to the varactor diode.
This transmitter is typical of small "traditional technology" VHF FM transmitters. More power is obtained simply by using the transmitter output to drive a power amplifier delivering the desired power. Because the output of an FM transmitter is constant in amplitude, a simple class C power amplifier can be used because linearity is not required. Tuneup is simple, with all adjustments made for maximum drive to the next stage (the collector current of a stage can be monitored while its input net work is adjusted), and finally all adjustments peaked for maximum output into a 50-ohm load. Although it is straightforward and simple, it uses several discrete components, which means more manufacturing costs. The increased number of stages needed to reach the higher frequencies (450 and 900 MHz), as well as the complexity of tuneup, makes a simpler approach more attractive in some cases. A PLL system can eliminate the need for the oscillator and multiplier chain in many cases. The next circuit discussed illustrates this point.
FIG. 2 Simple Audio Amplifier for Narrowband FM Transmitters
It is possible to use a single IC chip to perform the functions of transmitter low level signal generation. The Motorola MC13176D IC is a 16-pin surface-mount device that uses the PLL approach to generate a signal in the UHF band. We discuss a flea power (1 mW) transmitter that produces an NBFM audio signal on the frequency of 446 MHz. This happens to be the national simplex (direct communication, no repeater being used) frequency for FM two-way amateur communications in the 70-cm band. Although 1 mW may be dismissed as useless other than as a demonstration, this is not necessarily true. First, antennas are small at this frequency, with a full quarter-wave whip being a little more than 6 inches long. Radiation efficiency is high, and the ease of providing an adequate ground plane (the transmitter case and batteries in many instances being adequate) allows an efficient antenna system com pared to lower VHF frequencies. A good portion of 1 mile can be covered with power levels of a few milliwatts because the bandwidth used (13 kHz) allows a sensitive receiver to be used (typically 0.15-0.2 microvolt for 20 dB quieting for modern designs). Therefore, a useful short-range transmitter can be constructed from a single IC chip, and the whole transmitter powered with a 3-volt supply, two AA or AAA cells being adequate because current drain is only about 30 mA. If more power is desired, the output of 1-2 mW can be fed into another RF amplifier stage, but we did not pursue this approach because the power output is adequate for short-range use.
This circuit can be powered by a supply of 1.8-5 volts; however, remember that the varactor diode controls the transmitter frequency and its bias should be constant. A regulator can be used to derive a constant bias if a varying supply is used to power the transmitter. The bias can be 2-4 volts, as needed.
The MC13176D unfortunately is not available in standard DIP packages, only in rather experimenter-unfriendly surface mount. It is a fact of life that many of the newer chips that offer much to the experimenter turn out to be almost unusable because of the mechanical difficulty in working with them with ordinary hand tools.
These devices are designed for automated assembly and large-volume use in com pact equipment. This type of equipment (cell phones, computer-related items, etc.) often has a short life cycle (1-3 years) because of rapid technical obsolescence, must be low in cost, and is generally not designed to be repaired down to the component level. Pin spacings of 0.050 or 0.030 used in packaging these chips, with 24 pins or more, can be difficult to work with by the typical experimenter; however, the 8-pin SO8 and the 16-pin SO16 are not as bad as they may appear. Several easily available 8- and 16-pin devices are useful for RF applications. A study of the pin layouts of these devices and a little thought about circuit board layout often results in a layout that can be feasible to build with hand tools and a little care. A few of the pins are often grounds and supply pins, and only a few peripheral components are needed.
Chip resistors and capacitors of 60-by-120-millimeter size are fairly easy to work with. Also, 1/10 watt resistors and small ceramic discs can be used. There is often no real need to use an ultra-compact layout anyway. The small packages and short leads that are obtainable with surface mount permit good UHF performance with fewer of the circuit difficulties often experienced with circuits constructed from larger throughhole components. So, the experimenter should not turn away from using these devices because useful items can often be constructed. Examining the MC13176 data sheet ( FIG. 3) reveals that it is basically a PLL system with a VCO (in this case, actually a current-controlled oscillator, or CCO), divider (fixed at ÷32) reference oscillator, and phase detector. The reference oscillator is generally used with an external crystal, and the PLL output frequency is therefore 32 times the crystal frequency. This would be the same as that obtained using five doubler stages in a conventional multiplier chain. The PLL can be operated as high as 950 MHz. A balanced RF output is provided, with capability for AM or FM, or pulse modulation.
With a 3-volt supply, 1-2 mW output into a 50-ohm load is possible. Although it is possible to introduce audio into the PLL directly, NBFM of 5-kHz deviation is easily done with a varactor diode in series with the crystal. Because the output frequency is the crystal frequency times 32, only 150-Hz deviation is needed. With the required 13.9375-MHz fundamental cut crystal needed for 446.000-MHz output, this is easy to get, with good modulation linearity.
Motorola MC13176 UHF FM/ AM Transmitter Chip [Data Sheet]
FIG. 4 Schematic for a 446 MHz Amateur FM Transmitter
Examining the circuit of the transmitter ( FIG. 4), some rearrangement from the recommended (data sheet) circuit was made. In order to be more compatible with other circuits in this book, a negative ground supply is used. Good bypassing of the positive rail is obtained by chip capacitors C2, C3, C5, C9, and tantalum capacitor C4. Note that pins 2, 3, 5, 10, and 15 are grounded. Instead of expensive, difficult-to-find manufactured coils and chokes (for the home hobbyist without OEM accounts at several distributors), coils were wound from magnet wire. A simple twisted-wire balun was used at the output instead of the short length of rigid coaxial cable specified. This worked just as well, and the hobbyist can make it. L4 is a 1.5-3.5 (or more) microhenry slug-tuned coil ,and materials for this can be found in a junked TV or CB radio of the 1970s-1980s vintage. It consists of 19 turns of #32 gauge wire close-wound on a 1/4-inch-diameter slug tuned form. Moving the slug in and out varies the inductance. If preferred, a commercially made coil of suitable inductance can be used. The circuit works as follows:
Crystal X1, a 13.9375-MHz fundamental cut crystal, is used in conjunction with feedback capacitors C6 and C7 (47 pf chips) to act as an oscillator. Inductor L4 and varactor diode D1 (an MV2107, 22 pf at 4 volts bias) appear as a reactance network in series with the crystal. R3 acts as an isolation resistor and with R4, returns the varactor to ground. The cathode of the varactor is returned to the 3-volt supply and is reverse biased by this amount. Audio is coupled via C8 to R4 and appears across R4, and the audio voltage is effectively superimposed on the varactor bias. This causes the capacitance to vary at an audio rate, effectively modulating the reactance and therefore the oscillator frequency. Slug-tuned coil L4 is used to adjust the crystal frequency, so the output frequency is exactly 446.000 MHz. About 0.6 volts peak-to peak is required for the full 5-kHz deviation. This audio is obtained from an audio amplifier and is preferably limited to this level by using a clipper and lowpass filter to remove speech components above 3000 Hz. Also, suitable preemphasis should be used. This is usually a simple 6dB per octave rising response from 300-3000 Hz and improves the signal-to-noise ratio at the receiver. The audio amplifier, limiter, and clipper are straightforward and are not discussed here. A suitable circuit is shown in FIG. 2, which is discussed later. Limiting is obtained by driving the amplifier into saturation and cutoff because it operates from a 5-volt supply.
L1 is the current-controlled oscillator (CCO) coil. It is seven turns of #22 bare wire wound in the threads of an 8-32 screw as a mandrel. The screw is withdrawn after winding the coil, leaving an airwound solenoidal coil. This coil is slightly high in inductance, and in the circuit will have to have its turns spread somewhat to achieve the correct CCO frequency (446 MHz). The rest of the CCO components are inside the chip. A current from current amplifier Q1 controls the CCO. Loop filter components that determine loop characteristics are R5, R7, and C10. The output of the internal phase detector appears at pin 7. R6, R8, R9, and R10 are biasing resistors for amplifier Q1. The current into pin 6 controls the VCO frequency; it can be either a positive or a negative current. The loop amplifier Q1 supplements the 50 microampere current available from the internal source in the chip and provides a 100-microampere boost, increasing the available current swing and therefore the hold in range of the loop. RF output appears at the differential output pins 13 and 14 and is converted to single-ended output by L2. L2 is 2 inches of a twisted pair of #32 magnet wire and acts as a balun transformer. This is not critical, and two wires twisted about 10 twists per inch were used. C1 serves as a DC blocking capacitor.
RF choke L3 is nine turns of #24 magnet wire wound into an aircore solenoidal coil using an 8-32 threaded screw as a mandrel; it is not critical.
The current into pin 16 controls the RF output and is obtained via R1. About 2 mA is needed for maximum output. R1 could be made variable (use a 5K or 10K potentiometer) if you want to vary the output power. Pin 11 is a chip-enable pin and can be used as an on-off control. When pin 11 is open, the transmitter is powered down (off state).
This transmitter was breadboarded using a ready-made PC board from Motorola (see data sheet FIG. 3), and a layout similar to that shown should be used. If you are experienced in RF work, you can modify the layout as needed, but be sure to follow the suggestions given in the data sheet regarding proper bypassing and grounding. It worked the first time, and good results were obtained. Spurious outputs were measured as -35 dB down with respect to the carrier, which is not too bad considering the simplicity of the circuit. These spurs were spaced approximately 14 MHz from the carrier frequency and are caused by reference frequency mixing with the CO frequency. Proper filtering using tuned circuits can reduce them.
Using a pocket scanner, the transmitter signal was picked up loud and clear over 1/4 mile from the laboratory. The transmitter breadboard was sitting on the lab bench, feeding a 6-inch whip antenna. A 400-Hz audio tone was used for modulation, and the transmitter was set for 5-kHz deviation. A 3-volt power supply was used, and supply current was 28 mA. This is not bad performance at all. This circuit should find application wherever a low-power UHF audio transmitter is needed, and it makes for an interesting experimenter project. Note that a ham license is needed to place this transmitter on the air legally using the 446-MHz frequency.
An audio amplifier is needed for these FM transmitters. It should provide some limiting of output so as not to produce excessive frequency deviation. In addition, frequency response should be a 6 dB/octave rise to 3000 Hz, then a falloff of 12 or 18 dB/octave to eliminate excessive high frequencies and components generated during audio limiting. Audio limiting may be done with a clipper circuit using diodes or by designing the audio amplifier to limit by using a low-supply voltage to limit out put and making sure that the limiting is fairly symmetrical on positive and negative waveform excursions. A simple audio section suitable for use with these transmitters is shown in FIG. 2. A single transistor Q1 is biased by R1, R2, R3, and R4. A 5-volt regulated supply is used, and the bias point for the transistor is selected for simultaneous positive and negative peak limiting. About 4 volts peak-to-peak swing is obtained, and the gain of the stage is approximately 12 times at 1000 Hz. This allows a microphone or audio source of about 0.1 volt rms to drive the amplifier to full output. Coupling capacitor C3 couples audio to RC network R5, C4, R6, C5, R7, and C6. This network provides filtering of undesired high frequencies above 3000 Hz. The output of this network feeds a potentiometer R8, which is set so that the audio to the transmitter is sufficient to produce the required deviation (normally ±5 kHz), with maximum audio input level to the amplifier.
In addition, data in tone form (AFSK) may be fed into the amplifier if data is to be transmitted. Bandwidth is about 3000 Hz, which should handle any data that can be sent over a telephone line.
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