One of the most important parts in a radio receiver is the mixer. Fig. 1 A and B show an actual implementation as applied to our previous 9 MHz i-f communications receiver example. The level of the intermodulation performance of this device plays a crucial role in the design of such instrument, as we previously discussed. Today, the most commonly used mixer is the hot carrier diode type, although new technologies have recently evolved, as we will analyze later.
THE DIODE MIXER AND DIPLEXER
In using the diode mixer, one must pay particular attention to impedance matching particularly of the i-f output. Proper termination of this port helps achieve the predicted intermodulation distortion of the entire system. The use of a bandpass filter, called a diplexer is recommended for achieving this goal. Fig. 1C shows an actual implementation for the 9 MHz i-f communications receiver example. The diplexer terminates the unwanted products coming out of the mixer, sometimes up to the sixth order. Several diplexers can be used at the same time following a mixer in order to terminate different mixer products. Today there are other ways of matching the output of a mixer to the i-f amplifier as we will see later.
THE DOUBLY-BALANCED MIXER AND ITS PERFORMANCE CHARACTERISTICS
A doubly-balanced mixer works in the following manner: referring to Figs. 2 and 3 it can be seen that the voltage at the secondary of the local oscillator (LO) transformer forms currents that flow through the diode pair D1, D2, or D3, D4 according to polarity, causing alternate conduction, and making the ends of the secondary of the rf transformer points B and C to appear alternately at ground potential, at the frequency of the signal which is applied to the LO port.
Fig. 1. First mixer and diplexer.
Fig. 2. Electrical operation of a doubly-balanced mixer.
It can be seen that the signal appearing at the i-f output will be influenced by the level and polarity of the signal at the rf transformer's secondary, as well as by which terminal of that secondary is at ground potential at that time as shown in Fig. 2.
The output at the i-f port contains the sum and difference of the frequencies of the signals present at the LO and rf ports, plus other mixing products created by the same harmonic frequencies as discussed earlier. See Figs. 4 and 5.
Fig. 3 Output waveform resui:rig ff . orn toe switching action cf
a douby balanced mixer.
Fig. 4. Several doubly-balanced mixers (courtesy of Watkins-Johnson
Company).
Fig. 5. The WJ-M6E is an example of a Class I doubly-balanced mixer
(courtesy of Watkins-Johnson Company).
HARMONIC INTERMODULATION OF MIXERS
This type of intermodulation distortion is sometimes referred to as the spurious response of a diode mixer, and is created by the i -V transfer characteristics of the diode junctions involved, which generate all nF(1.0) ± mF(ro (where n and m = 0, 1, 2, 3 ... x) harmonic products. An important property of the doubly-balanced mixer is the suppression by common-mode cancellation of all internally-generated harmonic outputs which result from even-order harmonic components (Although odd harmonic components of the 3±3, 5±5 order cannot be suppressed by any mixer cancellation, they can be reduced by higher LO drives in Class II and Class III devices, as will be discussed later in the two-tone intermodulation ratio section.) of the LO and/or rf input signals, such as 2F(1.0) , 4F0.0) , 2Fuo , 2F(Lo) ± F. , F tro oLo 2FUt AND 2F(LO) ± 2F
This cancellation is, however, not a perfect matter due to the imbalance of the elements involved, even though great care is usually exercised in matching all diodes. For those interested, Fig. 6 shows suppression performance that can be expected from a Class I doubly-balanced mixer. There are 3 classes of doubly-balanced mixers, defined by the LO drive requirements:
Class I is + 7 dBm to + 9 dBm, Class H is + 17 dBm to + 20 dBm, and Class III + 23 dBm to + 27 dBm.
Fig. 6. Typical spurious response chart for Class I doubly-balanced
mixer.
AMPLITUDE MODULATION NOISE SUPPRESSION
Another property of the doubly-balanced mixer is the suppression of AM noise associated with the local oscillator. AM noise components on the other side of a local oscillator carrier will mix with the rf signal and produce noise sidebands within the i-f pass-band as we'll discuss later. While this phenomenon is common to all mixers, the balanced mixer tends to cancel this effect so that the noise sidebands in the i-f are typically down from the converted carrier level by 40 to 50 dB at frequencies up to 100 MHz. (The spurious levels will typically increase at higher frequencies.) It can be seen that the amount of AM noise suppression can actually determine the mixer's noise figure with a particularly noisy local oscillator. Put another way, it can determine the maximum local oscillator drive level allowed for the particular mixer for a desired noise figure. In spite of this, not all AM noise is cancelled in a DBM. Un-cancelled AM noise is sometimes caused at the i-f port, by the FM noise of the local oscillator moving through the slopes of the following i-f filter. AM noise suppression plays an important part in choosing a good doubly-balanced mixer and implementing it properly in a particular receiver design. The mechanization of such a test is, however, beyond the scope of this guide. The reader is again directed to the specialized material provided by mixer manufacturers. See Figs. 7 and 8 and Table 1.
Fig. 7. The Watkins-Johnson WJ-8888 B is a highly versatile hf receiver
which covers the 0.5 to 30 MHz frequency range in 10 Hz increments. Low
intermodulation distortion and wide dynamic range performance are achieved
in this design (courtesy of Watkins-Johnson Company).
Fig. 8. Simplified block diagram of the WJ-8888 receiver.
Table 1. Specifications of The W1-8888 Receiver.
CONVERSION LOSS IN DIODE MIXERS
Conversion loss in a diode mixer is usually referred to as the single sideband conversion loss, and is simply the power difference (expressed in dB) between the input rf level (expressed in dBm) and the output i-f level (expressed in dBm) of the desired product (sideband) as shown in Equation 1, where L c is the conversion loss.
P Lc(dB) = P - (rOdBm 6-f )dBm [Eq. 1]
If a - 10 dBm number is used for the rf input, and the desired sideband measures -16 dBm*, then:
L c = - 10 - (- 16) = 6 dB
Although single sideband conversion loss is usually specified by the mixer manufacturer, (typically 6 dB for a doubly-balanced mixer) its importance is crucial to the design of a radio receiver. Knowledge of this value dictates not only the amount of amplification needed to make up for it, but more importantly, it determines the noise figure of the mixer which impacts the total performance of a radio receiver as we will see in the next paragraph.
NOISE FIGURE OF MIXERS
Referred to as the single sideband noise figure, this value is a direct function of the conversion loss plus the estimated contribution in electrical noise caused by the diode junctions, a value which is typically 0.5 to 1 dB as shown in Equation 2.
SSB NF LC + 1 dB [Eq. 2]
As we discussed earlier in this guide, this number can determine the noise figure of the system, (if predominant) as well as the bottom level of the dynamic range of a particular stage and enters the total dynamic range picture in a multi-conversion receiver.
If a receiver were to be designed that has an expected noise contribution at the antenna of less than this number, such as in a space communication receiver, a lower noise figure mixer of a different type (or a lower noise figure preamplifier used ahead of the doubly-balanced mixer) should be used in order to bring down the bottom level of the dynamic range.
TWO-TONE INTERMODULATION RATIO
The intercept method described earlier for the overall performance of a radio receiver is a form of a two-tone intermodulation ratio test which can be applied to all or any parts that make up the system, including amplifiers (rf, i-f and af) and mixers. The total performance depends on the individual performance of these parts, and the mixer is often the most important contributing element to this performance. Shown in Fig. 9 is a typical two-tone test setup.
[This experiment requires a spectrum analyzer. or a power meter. ]
Fig. 9. Typical two-tone test set-up.
When two F td, signals are simultaneously applied to a mixer they will combine with the F(Lo) producing the sums and the differences as well as the undesired intermodulation products, as discussed earlier. The largest magnitude of undesired intermodulation product level is the third-order product and it was chosen as a means of performance specification for mixers as well. The Two-Tone Intermodulation ration is the ratio of the third order IM product to an i-f output level, at a specified power level for the two rf inputs.
Fig. 10. Two-tone intermodulation distortion for doubly-balanced
mixers.
Fig. 11. Internal view of the WJ-M9G mixer (courtesy of Watkins-Johnson
Company).
If the output products versus rf input power are plotted on a log-log graph, the third order IM products present a 3:1 slope while the i-f outputs present a 1:1 slope. The two slopes meet at the third order intercept point as shown in Fig. 10. The intercept point is specified in terms of its rf input power level and is the means of expressing the mixer's performance. See Fig. 11.
A rule of thumb is that the higher the intercept point, the better the mixer performance. Generally, improved performance can be obtained with higher LO drive levels which will switch the diodes further into the linear regions at both ends of their I-V curves while reducing the amount of time spent in the nonlinear region centered around V=0. While the forward current through each diode pair is almost unlimited, the reverse voltage is limited by the forward voltage drop of the un-conducting diode pair.
The solution for this problem is adding more series diodes, adding series resistance to each diode, or choosing diodes having a higher break down voltage. This in turn increases the requirement for LO drive levels, which increases power consumption. The improvement in intercept point as effected by the local oscillator power is exemplified in the test results shown in Fig. 12. In this experiment, performed by Watkins-Johnson Co., three typical classes of mixers were tested in a two-tone intermodulation setup. The results show clearly that the higher the class is, (and consequently the higher LO drive requirement) the better the intercept point is. For example: the class I, WJ-M62 mixer requires + 9 dBm of LO drive power and has an intercept point of + 13 dBm, while the class II, WJ-M40 mixer with a requirement of + 20 dBm LO drive power, exhibits a superior intercept point of + 20 dBm. The best intercept point in this experiment was obtained with a class III mixer, the WJ-M9E which has an intercept point of + 33 dBm, using LO power of + 23 dBm.
Fig. 12. Typical two tone intermodulation performance for 3 classes
of mixers, showing improvement in performance with Increased LO drive
(courtesy of Watkins-Johnson Company).
Spectrum analyzer photographs for the intermodulation performance of 3 classes of mixers are shown in Fig. 13 A, B, and C. They also show reduced intermodulation distortion with increased LO drive for the typical 3 classes of mixers, in this case a WJ-M1 for class I, a WJ-M9BC for the class II, and the WJ-M9E for the class III.
COMPRESSION POINT (-1 dB)
When a single rf input is applied to the rf port of a mixer, along with a large LO signal, the conduction transfer characteristic of the diodes changes as the rf input level is increased beyond a point, and the output will eventually start saturating until no increase is obtained no matter how much the rf input level is increased.
The point at which the output level drops 1 dB from following the input is the -1 dB compression point (Fig. 14). This point plays an important role in determining the upper limit of the dynamic range of a radio receiver.
DESENSITIZATION LEVEL
Fig. 13. Spectrum analyzer representation for the intermodulation
performance of 3 Classes of mixers (courtesy of Watkins-Johnson Company).
Good performance is achieved with a Class I mixer, the WJ-M1. LO= +7
dBm (A). Better performance is achieved with a Class II mixer, the WJ-M9BC.
LO= +17 dBm (B). Best performance is achieved with a Class Ill mixer,
the WJ-M9E. LO= +27 dBm (C).
Fig. 13.
(Continued)
Another receiver parameter that depends on the mixer performance is the 1 dB desensitization created by a mixer. This level is the rf input power of an interfering signal that causes the small-signal conversion loss to increase by 1 dB. This level is typically 2 to 5 dB above the compression point level and should not be confused with it. Fig. 15 shows a typical setup for measuring this parameter.
ISOLATION IN BALANCED DIODE MIXERS
Isolation between two mixer ports is expressed by the amount of attenuation that an input signal at one port experiences when measured at the other port. This parameter is usually important in preventing unwanted signals from getting back into the system.
Fig. 14. Finding the compression point.
Fig. 15. Typical measurement set-up for the desensitization level
of a doubly-balanced mixer.
Although most doubly-balanced mixers have good isolation (typ. 50 dB) some systems might require additional filters and amplifiers for improving this characteristic. The setups shown in Fig. 16 A and B are used to measure this isolation. See Fig. 17.
Fig. 16. Isolation test set-up. LO to if isolation test (A). Ftf
to i-f isolation test (B).
Fig. 17. Internal construction of the WJ-M90 high level doubly-balanced
mixer (courtesy of Watkins-Johnson Company).
OTHER TYPES OF DIODE MIXERS
While the doubly-balanced diode mixer remains the most popular type, there are other types in use today.
THE SINGLY-BALANCED MIXER
Fig. 18 shows a singly-balanced mixer which uses a balanced transformer to feed the LO signal to the two matched mixing diodes, while the rf signal is coupled via a high-pass LC network. If the circuit is truly balanced and symmetric, the voltage dropped in each diode is the same and consequently, no LO signal will appear at the rf port and vice-versa.
Filtering is generally used in the i-f port in order to increase the isolation between the rf and the i-f port as shown.
Fig. 18. Representative schematic of a singly-balanced mixer.
Fig. 19. Internal configuration tor an image-rejection mixer.
IMAGE-REJECTION MIXERS
The image problem discussed earlier in receivers with relatively low i-fs can be improved with the help of the so-called image rejection mixers.
This is particularly true in guidance receivers which, because of their tight packaging requirements cannot afford double-conversion approaches or preselectors. Doubly-balanced mixers are combined with other wideband devices to accomplish this type of mixer. Fig. 19 shows a pair of gain-and phase-matched mixers arranged to provide image rejection by cancellation in the hybrids. A low-i-f radio receiver employing such a mixer will exhibit a typical image rejection of 20 to 30 dB without any pre-selection.
IMAGE-RECOVERY MIXERS
We previously discussed how mixers respond to signals at the image frequency, however mixers can also generate energy at this frequency.
This phenomenon is accomplished through two mechanisms. First, the second harmonic of the LO can mix with the incoming rf signal creating energy at the image frequency as shown in Equation 3 = 2FLO F rf
Eq. 3 F_image (internally generated)
Secondly image energy can also be created by the i-f signal being reflected back into the mixer (do to mismatching at the i-f port) and remixing with the LO energy as shown in Equation 4.
Eq. 4 F_image (internally generated) = F LO ± Fi-f
To verify these mechanisms, the reader is encouraged to use the image model previously presented in Fig. 4-7. If using the values from the example in Equations 3 and 4 we can easily prove the internal generation of image energy in mixers. In the case of Equation 3:
= (2 x 14.545) - 14.090 = 15.000 MHz. F_image (internally generated)
In the case of Equation 4:
F. = 14.545 + 455 = 15.000 MHz. image (internally ¡tene men)
Fig. 20. The image-recovery mixer principle.
In both cases, the image created energy can be recovered and used so as to create additional power at the i-f frequency, reducing the conversion loss of the mixer. Fig. 20 shows the implementation of an image recovery mixer.
In this arrangement the filters at the rf and LO ports of the mixer serve the function of reflecting the internally generated image energy, back into the mixer, while providing a path for the rf and the LO signals. Proper adjustment of the electrical lengths e and e' between the filters and the mixer provides the right phasing between the two resultant i-f contributions due to signal and image. This in turn results in a decrease in the conversion loss of the mixer. Typical improvements of 2 dB have been reported with this type of mixer.
Although not widely used, the image recovery mixer finds increased application in systems where gain cost is at a premium. It is important to note, however, that if not properly designed, the image recovery mixer can be lossier than a typical balanced mixer, presenting a problem rather than a solution to a communications system.
JFET MIXERS
JFETs are also used and are gaining a good intermodulation reputation. Fig. 21 shows a favorite scheme, with a carefully balanced and matched set of FETs which provides a +30 dBm typical intercept point.
Other FET mixers use VMOS devices. Shown in Fig. 22 is a suggested singly-balanced mixer using the Siliconix VMP 4 FET. With proper balance of the dc bias, a typical 43 dBm intercept point has been reported.*
'Modem Receiver Mixers for High Dynamic Range. Doug DeMaw and George Collins. PST. January. 1981.
Fig. 21. Doubly-balanced mixer using tour closely-matched junction
FETS.
Fig. 22. Suggested singly-balanced mixer using the Silicon bc VMP 4 V-MOS FET.
INTEGRATED CIRCUIT MIXERS
Active mixers in IC form are also common. They are inexpensive and provide gain along with mixing, an important factor when designing economically. Fig. 23 shows the internal configuration of a motorola MC1596. Also shown is the surrounding circuitry required to achieve an active doubly-balanced mixer with this IC.
IMPLEMENTATION OF IC MIXERS IN BILATERAL SYSTEMS
Fig. 23. Internal configuration and application circuit for an MC1596
active doubly-balanced mixer.
Fig. 24. Implementation of active mixers in a bilateral transceiver.
Fig. 25. The heart of a bilateral transceiver using MC 1596 active
mixers, Stoner-Goral model 711 (courtesy of SGC Inc.).
Fig. 26. The Stoner-Goral model SG-710 uses an improved bilateral
system (courtesy of SGC Inc.).
Table 2. Trade-Offs in Mixer Technologies.
Active mixers are widely used in systems like the transceiver shown in Fig. 24. This scheme is economical to build, and is ingenious for achieving a transceiver with a minimum number of parts. There are two MC1596s in this system. The antenna is directly coupled to the first one, and the first oscillator provides conversion with gain (about 50 dB) to a 10.7 MHz i-f. The second oscillator (bfo) injects into a second MC1596 and provides detection with another 50 dB of gain. The MC1596 can be used either as a first mixer, a product detector, or a sideband generator, with only minor changes in values. This system takes advantage of this fact by using a switching mechanism involving relays and diodes transforming the receiver into a transmitter. The actual implementation of the heart of this transceiver is shown in Fig. 25. An improved version of this approach is used in the transceiver shown in Fig. 26 from Stoner-Goral Corporation.
MIXER TECHNOLOGY TRADE-OFFS
It has been seen that linearization of mixer performance plays an important role in the design of a communication receiver. Several types of mixers were described and a few technologies were analyzed. Today, the doubly-balanced diode mixer is the most widely used, and despite its relatively high conversion loss (6 dB typical) it provides a typical dynamic range approaching 100 dB with high output intercepts (>+23 dBm). Its disadvantages are that it is susceptible to odd-order harmonic mixing and it requires a proper termination.
New recent developments in doubly-balanced mixer design indicate third-order intercept point achievements of + 38 dBm (VAY-1, from Mini Circuits Labs.) with LO drives in excess of 28 dBm. The JFET mixer, while relatively more expensive, provides superior noise figures (4 dB typical), combined with gain and high intercept (> + 25 dBm). The bipolar transistor in an active doubly-balanced form will continue to provide a low-cost approach, while experiments with MOS and VMOS FETS look promising from a performance standpoint. Shown in Table 2 is a list of trade-offs between some of the rf mixer technologies.